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Switched Reluctance Machine has progressed as a suitable contender in the family of electrical machines for its high starting torque and better performance in different operating conditions. The need of accurate modeling of this machine is therefore dependent on the accurate determination of losses of which iron loss is exigent. This paper aims to determine iron loss from the equivalent circuit of the machine based on the input data tables that are directly obtained from the experiments performed on an existing 4-phase switched reluctance machine. The experimental determination of the iron losses has been compared with those who were predicted under the same operating conditions.&lt;br/&gt;&lt;br/&gt;Equivalent Circuit of Switched Reluctance Motor&lt;br/&gt;&lt;br/&gt;The iron losses calculation from the equivalent circuit of the switched reluctance machine (SRM) is presented in [1] that included two ends i.e., unaligned and miss-aligned positions of the machine. Whereas an intermediate position was not taken into consideration. Fig. 1 shows an equivalent circuit in which R represents phase winding resistance, &lt;i&gt;L&lt;/i&gt; is self-inductance, and resistive branch ‘&lt;i&gt;r’&lt;/i&gt; that is dependent on rotor position, change in flux linkage, and frequency of phase switching. This element is connected in parallel to ‘&lt;i&gt;L’&lt;/i&gt; to represent the iron loss in the core of the machine [1].&lt;br/&gt;The arrangement i.e., the parallel combination of ‘&lt;i&gt;r’ &lt;/i&gt;and ‘&lt;i&gt;L&lt;/i&gt;’ shown in the circuit Fig. 1 [1] describes the flux penetration and eddy current in the magnetic core of the motor. A review on different methods for loss calculations of SRM is given in [2].&lt;br/&gt;&lt;br/&gt;Results&lt;br/&gt;&lt;br/&gt;For purpose of the illustration, the waveforms of currents and in the equivalent circuit are shown in Fig.2 (a). Fig.2 (b) shows the equivalent iron-loss-resistance vs. flux- linkage in the intermediate position of currents and in the equivalent circuit of Fig.1.

**References:**

1. Corda, J. and Jamil, S.M., 2009. Experimental determination of equivalent-circuit parameters of a tubular switched reluctance machine with solid-steel magnetic core. &lt;i&gt;IEEE Transactions on Industrial Electronics&lt;/i&gt;, &lt;i&gt;57&lt;/i&gt;(1), pp.304-310.&lt;br/&gt;2. Li, S., Zhang, S., Habetler, T.G., and Harley, R.G., 2019. Modeling, Design Optimization, and Applications of Switched Reluctance Machines—A Review. &lt;i&gt;IEEE Transactions on Industry Applications&lt;/i&gt;, &lt;i&gt;55&lt;/i&gt;(3), pp.2660-2681.

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INTERMAG 2021

 Experimental determination of equivalent iron loss resistance for prediction of iron losses in a Switched Reluctance machine

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Switched Reluctance Machine has progressed as a suitable contender in the family of electrical machines for its high starting torque and better performance in different operating conditions. The need of accurate modeling of this machine is therefore dependent on the accurate determination of losses of which iron loss is exigent. This paper aims to determine iron loss from the equivalent circuit of the machine based on the input data tables that are directly obtained from the experiments performed on an existing 4-phase switched reluctance machine. The experimental determination of the iron losses has been compared with those who were predicted under the same operating conditions. Equivalent Circuit of Switched Reluctance Motor The iron losses calculation from the equivalent circuit of the switched reluctance machine (SRM) is presented in [1] that included two ends i.e., unaligned and miss-aligned positions of the machine. Whereas an intermediate position was not taken into consideration. Fig. 1 shows an equivalent circuit in which R represents phase winding resistance, L is self-inductance, and resistive branch ‘r’ that is dependent on rotor position, change in flux linkage, and frequency of phase switching. This element is connected in parallel to ‘L’ to represent the iron loss in the core of the machine [1]. The arrangement i.e., the parallel combination of ‘r’ and ‘L’ shown in the circuit Fig. 1 [1] describes the flux penetration and eddy current in the magnetic core of the motor. A review on different methods for loss calculations of SRM is given in [2]. Results For purpose of the illustration, the waveforms of currents and in the equivalent circuit are shown in Fig.2 (a). Fig.2 (b) shows the equivalent iron-loss-resistance vs. flux- linkage in the intermediate position of currents and in the equivalent circuit of Fig.1.

**References:**

1. Corda, J. and Jamil, S.M., 2009. Experimental determination of equivalent-circuit parameters of a tubular switched reluctance machine with solid-steel magnetic core. IEEE Transactions on Industrial Electronics, 57(1), pp.304-310. 2. Li, S., Zhang, S., Habetler, T.G., and Harley, R.G., 2019. Modeling, Design Optimization, and Applications of Switched Reluctance Machines—A Review. IEEE Transactions on Industry Applications, 55(3), pp.2660-2681.

**Images:**
![](https://assets.underline.io/uploads/markdown_image/1/image/5294b7a5d713fb5088771a6e3f541dca.jpg)

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poster

I. INTRODUCTION Nowadays, the demands of high torque/power rating in critical industrial applications usually lead to bulky size and excessive material consumption in the conventional permanent magnet (PM) machines [1]. Thus, torque density improvement is widely recognized as a challenging issue for traditional PM machines [1]. As a common practice, doubling the stator is normally employed to increase the torque density within a limited overall dimension, namely, the dual-stator (DS) machines [2]-[4]. For the DS machines, the thermal management and protection issues associated with the PMs tend to be problematic issues with the cantilever cupped rotor. The stator-PM DS machines [4] can well facilitate the PM heat dissipation, but also have a simple and robust rotor which is preferable for high-speed critical-safety operation. Nevertheless, the inner and outer stators in the preceding stator-PM DS structures are structurally duplicated, the geometric conflicts between the PM excitations and windings on the stator still exist similar to those single stator counterparts. Therefore, this paper aims to propose novel DS biased PM machines (DS-BPMM) only having biased PM excitations in the inner stator. Different from the existing DS machines having two identical stators, the developed DS-BPMMs combines the advantages of the design concept of “partitioned stator (PS) [5]” and the conventional DS machines. In the perspective of the geometry, the proposed machines can be considered as the synergies of an outer-rotor stator PM machine plus a separate outer stator similar to traditional fractional-slot machines. As a result, the space tradeoff between PM and copper areas, as well as severe magnetic saturation are significantly alleviated compared to the traditional DS counterparts. In this paper, the electromagnetic characteristics of the DS-BPMMs with different PM arrangements in the inner stator are investigated and compared. The machine topology and operating principle based on interior PM (IPM) and surface-mounted PM (SPM) structures will be addressed. Finally, an IPM-DS-BPMM prototype is manufactured and tested to verify the theoretical analyses. II. MACHINE TOPOLOGIES AND STATOR/ROTOR POLE COMBINATION Figs. 1 (a) and (b) show the proposed machine topologies, which can be regarded as IPM and SPM types, respectively. It should be noted that the total PM usage, outer stator, stack length, and rated current are kept the same for the two designs to offer a fair comparison. Meanwhile, the two machines are both characterized by a 6/11/6 outer-stator-pole/rotor-segment/inner-stator-pole configuration. Since the two machines comply with the variable reluctance principle, their stator/rotor pole combinations can be governed by [6] Ns/GCD(Ns, Nr)=km (1) where GCD signifies the Greatest Common Divisor between the stator pole number Ns and rotor pole number Nr, m is the number of phases, and k is a positive integer. It should be emphasized that the outer stator pole number is identical to that of the inner stator pole in these two machines. Besides, the flux linkage of single-coil is bipolar for SF machine, while unipolar for VR machine. However, the resultant phase flux linkage of VR topology is bipolar, which is mainly attributed to the opposite connection of armature windings in the same phase, as evidenced by the coil and phase flux linkages in the outer stator windings in Figs. 1 and (c) and (d), respectively. III. ELECTROMAGNETIC PERFORMANCE COMPARISON The open-circuit back-EMFs of the two machines are shown in Fig. 2(a). The significantly high EMF magnitudes can be observed in the IPM case. This is mainly attributed to the higher resultant phase flux linkage of the IPM machine due to its bipolar coil flux linkage. Accordingly, the SPM machine exhibits relatively lower torque capability compared to the IPM counterpart as reflected in the rated-load steady torque in Fig. 2(b). Meanwhile, it can be observed that the IPM one shows ~higher torque ripple ratio than the SPM one, which is undesired for the applications requiring smooth torque performance. IV. EXPERIMENTAL VALIDATION Based on the analysis abovementioned, the IPM configuration is selected and manufactured as shown in Fig. 2(c). The finite element (FE) and measured back-EMF waveforms are shown in Fig. 2(d), in which the good agreement between FE predictions and test results are achieved. The detailed characteristics analysis, comparison results, and experimental validation will be presented in the full paper.

**References:**

[1] G. A. Capolino and A. Cavagnino, “New trends in electrical machines technology,” IEEE Trans. Ind. Electron., vol. 61, no. 8, pp. 4281-4285, Aug. 2014. [2] M. Abbasian, M. Moallem, and B. Fahimi, “Double-stator switched reluctance machines (DSSRM): fundamentals and magnetic force analysis,” IEEE Trans. Energy Convers., vol. 25, no. 3, pp. 589-597, Sep. 2010. [3] S. Niu, K. T. Chau, J. Z. Jiang, and C. Liu. “Design and control of a new double-stator cup-rotor permanent-magnet machine for wind power generation,” IEEE Trans. Magn., vol. 43, no. 6, pp. 2501-2503, Jun. 2007. [4] D. Kim, H. Hwang, S. Bae, and C. Lee.” Analysis and design of a double-stator flux-switching permanent magnet machine using ferrite magnet in hybrid electric vehicles,” IEEE Trans. Magn., vo. 52, no. 7, pp. 1-4, Jul. 2016. [5] D. Evans and Z. Q. Zhu, “Novel partitioned stator switched flux permanent magnet machines,” IEEE Trans. Magn., vol. 51, no. 1, Art. no. 8100114, Jan. 2015. [6] J. T. Chen, and Z. Q. Zhu, “Winding configurations and optimal stator and rotor pole combination of flux-switching PM brushless AC machines,” IEEE Trans. Energy Convers., vol. 25, no. 2, pp. 293 - 302, Jun. 2010.

**Images:**

![](https://assets.underline.io/uploads/markdown_image/1/image/aedc4c0d896721af698b5502619d7a66.jpg)
Fig. 1. Topologies and flux linkage waveforms of the DS-BPMMs.

![](https://assets.underline.io/uploads/markdown_image/1/image/2ea5c842c25e8e95ad6c73004ee52f0f.jpg)
Fig. 2. Electromagnetic performance comparison and test result. (a) Back-EMFs. (b) Rated-load steady torque waveforms. (c) Prototype. (d). FE predicted and measured phase EMF waveforms.

 Comparative Study of Novel Dual Stator Machines Having Different Biased PM Configurations

I.Introduction The inter-turn short-circuit (ITSC) faults can cause excessively high short-circuit current in PM motors. This elevated current level will increase the temperature of the faulty windings and further melt the insulation of the nearby turns. As a result, some more serious faults can be induced, such as the interphase short-circuit fault, phase-to-ground short-circuit fault, and PM demagnetization [1]. The axial split phase permanent magnet synchronous motor (ASP-PMSM) has better isolation performance between phases compared to traditional circumferential split phase permanent magnet synchronous motor (CSP-PMSM) [2], which is benefit in preventing the spread of fault between phases. In this paper the behavior difference of ITSC fault between ASP-PMSM and CSP-PMSM has been investigated. Through theoretical analysis, finite element analysis (FEA) and experimental verification, it is found that compared to the CSP-PMSM, the ASP-PMSM has inherently advantages in mitigating the severity of ITSC fault. II.Modeling And Analysis of ITSC Fault A. Topology and advantages. As shown in Fig.1 (a), the phase-stators of ASP-PMSM are independently distributed along axial direction, which realized excellent magnetic, physical and thermal isolation between phases. These features are benefit in preventing the spread of fault between phases. There is only one phase in the circumferential direction of ASP-PMSM, and therefore the number of turn in series per phase of the ASP-PMSM is closed to three times that of CSP-PMSM. B. The behavior of fault current. This section build the electrical model of PMSM with ITSC fault, as is shown in Fig.1(b). For an incipient fault, the contact resistance is not ideal zero. Therefore, the contact resister rf is included in this model to consider the degradation process of insulation materials. Based on this model, the expression of fault current if is derived. |if|=|Vf|/(Rf+Rs+jωLaNs/N1) Vf=ea-Rf+jωLa(N1-Ns)/N1ia Rf=Nrf/Ns In above equation, N, N1 and Ns represent number of turns in series, number of turns per coil and number of shorted turns; respectively; and represent the phase back-EMF vector and current vector, respectively; La represents the self-inductance of phase A; Rs represents the phase resistance. The vector diagram of Vf for CSP-PMSM and ASP-PMSM is shown in Fig.1 (c) and (d), respectively, the blue line and black line represents the vector at low speed and high speed, respectively. Because the number of turns in series of ASP-PMSM is triple that of CSP-PMSM, therefore the Rf for ASP-PMSM is triple that of CSP-PMSM. At low speed, the amplitude of vector Vf-ls in Fig 1(d) is three times larger than that in Fig.1 (c). Thus the fault current of ASP-PMSM is larger than that of CSP-PMSM. However, for an incipient fault, the fault current of ASP-PMSM and CSP-PMSM are both small at low speed. At high speed, the amplitude of vector Vf-hs in Fig.1 (d) is smaller than that in Fig.1 (c). Thus the fault current of ASP-PMSM is smaller than that of CSP-PMSM. It can also be seen from Fig.1 (c) and (d) that the higher the fault resistance, the higher the speed to obtain the minimum short circuit current, that is, for an incipient fault, the ASP-PMSM can obtain lower fault current than CSP-PMSM at relative high operating speed. III.Simulation Analysis The fault current of ASP-PMSM and CSP-PMSM at different contact resistance and operating speed combinations is calculated by FEA and shown in Fig.1 (e) and (f). It can be seen that when the fault resistance is relatively large, the fault current of ASP-PMSM is small in the full speed range. This point is very benefit in limiting the further deterioration of fault and expanding the span life of the faulty motor. The simulation results are consistent with the theoretical analysis results in section II. The variation of fault current with the number of shorted turns at different contact resistance conditions is also investigated. The losses of two types of motors under healthy and different fault conditions are compared. It is verified that the copper loss of ASP-PMSM at different ITSC fault conditions is much lower than that of CSP-PMSM, which proved that the ASP-PMSM itself can mitigating the fault severity of ITSC fault compared to that of CSP-PMSM. IV.Experimental Test A 6.3kW prototype is manufactured, and the test platform is shown in Fig.2(a). The variation of one-turn short-circuit current with the operating speed at different contact resistance conditions is tested, and shown in Fig.2(b) and (c), which proved the accuracy of the above analysis. V.Conclusion 1) For an incipient fault, the ASP-PMSM can achieve small fault current at much wider speed range than that of CSP-PMSM, and the ASP-PMSM investigated in this paper has realized low fault current within full speed range. This is very useful to limit the further deterioration of fault and expanding the span life of the faulty motor. 2) With the same number of shorted turns, the loss of ASP-PMSM are much smaller than that of CSP-PMSM, it is proved that the ASP-PMSM itself can mitigating the fault severity of ITSC compared to that of CSP-PMSM. 3) The experimental test verified the accuracy of the theoretical analysis and the FEA.

**References:**

[1] N. Leboeuf, T. Boileau, B. Nahid-Mobarakeh, F. Meibody-Tabar, and G. Clerc, “Real-time detection of inter-turn faults in PM drives using back EMF estimation and residual analysis, ” IEEE Trans. Industry Applications, vol. 47, no. 6, pp. 2402-2412, 2011. [2] Feng Chai, Lina Geng, Yulong Pei,“A novel fault tolerant motor with integral slot non-overlapping concentrated Winding,” IEEE Access, vol.7, PP. 99462 – 99469, Jul. 2019.

**Images:**

![](https://assets.underline.io/uploads/markdown_image/1/image/d7bf3ea3d54fb2244b7850515ad0103a.jpg)
Fig.1 (a)Motor topology (b) Electrical circuit model. Vector diagram of (c) CSP-PMSM and (d) ASP-PMSM at different speed. The one-turn fault current varies with speed when rf is (e) 5mΩ and (f) 10mΩ.

![](https://assets.underline.io/uploads/markdown_image/1/image/b629ca0b5935825e7e8a30b58562c801.jpg)
Fig.2 (a) The test platform. The one-turn ITSC fault current varies with speed at (b) zero contact resistance condition and (b) non-zero contact resistance condition.

 Analysis of Inter-turn Short-Circuit Faults of Axial Split Phase Permanent Magnet Synchronous Motors

I. Introduction Direct drive wave energy conversion (DDWEC) device using permanent magnet linear generator (PMLG) eliminates the inefficient transmission device and achieves simpler mechanical structure. However, due to the direct connection between buoy and translator, the low-speed linear reciprocating wave motion will be directly transmitted to the translator. So PMLG needs to adopt a low-speed design [1]. It is necessary to increase the number of pole pairs of the generator or use power electronic devices to increase the output voltage and power density. But this will increase the loss and reduce the efficiency of the whole system. The linear-rotating axial flux permanent magnet generator (AFPMG) is proposed to improve the output voltage and power density of the DDWEC. By employing the magnet screw (MS) and AFPMG, it combines the merits of MS and AFPMG and realizes the soft connection between the buoy and generator rotor. The nut and screw of MS are directly connected to the buoy and generator rotor, respectively. When approximately 0.5m/s reciprocating wave motion comes, the movement can be accelerated by MS and magnetic field modulation and finally the speed of the rotor will reach 1200 rpm/min, which can improve the output voltage and power density consequently. While this structure brings advantages, at the same time, due to the use of a large number of rare-earth PMs, it will increase power generation costs and reduce practical competitiveness. Therefore, a hybrid excitation axial flux permanent magnet generator (HE-AFPMG) is proposed to reduce the amount of PMs and become a candidate for DDWEC. Not only that, because the wave conditions (typically 3-20s) are constantly changing, the corresponding wave excitation forces of the same buoy vary widely. By using hybrid excitation, the excitation current can easily adjusted to regulate the magnetic field of the air gap [2], which make generator suitable for different working conditions in wave energy conversion. II. Machine Configuration and Operation Principle In the proposed topology, as shown in Fig. 1, the armature winding and field winding are placed in upper stator and lower stator of AFPMG, respectively. PMs are placed at the slots of the upper stator and work as parallel excitation source. There is no risk of irreversible demagnetization in PMs with this configuration, meanwhile, it can reduce flux linkage and back electromotive force (EMF) harmonics. The operation principle of AFPMG based on modulation effect is similar to a magnetic gear and a magnetically geared machine when the rotor is consists of modulation teeth. The magnetic field will be accelerated due to the magnetic gear effect, when the excitation magnetic field passes through the air gap and modulation teeth. Eventually, potentials are induced in the armature windings. The number of pole pairs and speed of armature magnetic field are expressed as follow: paw = | pfw ± pmt | Ωaw = ( pfwΩfw ± pmtΩmt ) / ( pfw ± pmt ) where paw, Ωaw, pfw, Ωfw are the number of pole pairs and rotational speed of magnetic field in armature winding and field winding, respectively. pmt and Ωmt are the number and rotational speed of modulation teeth, respectively. In proposed AFPMG, the upper stator pole number Nus = 12, hence the number of pole pairs of magnetic field in armature winding paw = 2 or paw = 1 when the number of modulation teeth satisfies pmt = Nus ±2 or pmt = Nus ±1. Fig.1 III. Results Fig.2 The output power versus excitation current in proposed HE-AFPMG with armature winding having different load current is shown in Fig.2. IV. Conclusion The application of hybrid excitation in AFPMG used in DDWEC can eliminate risk of irreversible demagnetization in PMs and reduce flux linkage and back electromotive force harmonics. Meanwhile, it can adjust the air gap magnetic field to suit different sea conditions.

**References:**

[1] W. Li, K. T. Chau and J. Z. Jiang, "Application of Linear Magnetic Gears for Pseudo-Direct-Drive Oceanic Wave Energy Harvesting," in IEEE Transactions on Magnetics, vol. 47, no. 10, pp. 2624-2627, Oct. 2011. [2] C. Ye, Y. Du, J. Yang, X. Liang, F. Xiong and W. Xu, "Research of an Axial Flux Stator Partition Hybrid Excitation Brushless Synchronous Generator," in IEEE Transactions on Magnetics, vol. 54, no. 11, pp. 1-4, Nov. 2018.

**Images:**

![](https://assets.underline.io/uploads/markdown_image/1/image/951d474bb6205f855448b547badaaa7c.jpg)
Fig. 1

![](https://assets.underline.io/uploads/markdown_image/1/image/53fb22268e717a1e38e511372a0a531b.jpg)
Fig. 2

 A Hybrid Excitation Axial Flux Permanent Magnet Generator for Direct Drive Wave Energy Conversion

Due to the widespread concern of the greenhouse effect, people's requirements for energy saving and emission reduction of drive systems are becoming higher. The electrification of drive systems seems to be an irresistible trend. As the core of electric drive system, electric motor has been widely mentioned. The Permanent Magnet Synchronous Machine (PMSM) proved to be better in energy conversion which efficiency is higher than conventional Induction Machines [1-2]. In the application field of commercial logistics vehicles, the trend of electrification is increasing and the space occupied by the electric drive system is limited. As a result, a higher demand for density is required. Conventional electric motors applied in this platform are mainly Radial Flux Permanent Machines (RFPM). In particular, the application of the logistics EV traction motor requires low maximum speed but high output torque. The conventional design developed on this platform is a PMSM using traditional round copper wire as winding part and the rotor adopts “delta” shape topology. The “flux barrier” structure was added to increase the reluctance torque ratio, which was aimed to increase torque density of the whole system. However, due to the further reduction in already limited space, it will be difficult to meet the requirements using RFPM. Since the topological structure of the disc motor was proposed in last 70s [3-4], many researches have been done, especially on AFPM’s topological structures. Among them, some classic structures are: the TORUS (double-rotor-one-stator) structure, whose stator part is composed of copper winding and iron core, and the rotor part is composed of the surface mounted PMs and back iron; the AFIR (double-stator-one-rotor) structure, whose stator part is the same, but rotor part is composed only of the fan-shaped PMs [5-6]. As there is no back iron structure in the rotor part for AFIR, its rotational inertia is quite low and resulting the starting torque is low [7-8]. This advantage attracts us because the size requirements are more stringent for our platform. Based on this consideration, the optimal structure of the traction motor is selected as the AFIR structure with two stators and one rotor. In AFPM design, the rotor Permanent Magnets (PM) are directly in contact with the air gap, and the complex magnetic field has rich harmonic content, resulting in a higher proportion of eddy current losses in rotor PMs. Which can cause higher temperature rise in the rotor of an AFIR structure AFPM. Reducing PM eddy current loss is significantly important to the PM topology design, rotor temperature rise and overall performance. The AFPM in this paper adopts an AFIR 24s/20p design. The rotor PMs are segmented with identical radial distance, in order to reduce the length of the eddy current path in each PM and reduce the eddy-current loss, thereby reducing the risk of PM demagnetization caused by high temperature. In this paper the main body is divided in three parts, in part II, the design problems will be presented, which mainly includes: 1) Stator and rotor design, 2) The influence of the rotor support material on the motor, 3) The magnetic field measurement on the surface of the segmented magnet, 4) difference analysis between simulation and test. Part III will mainly introduce the problems encountered and the corresponding solutions during the manufacturing process, which includes: 1) assembly problem due to strong magnetic force, 2) air-gap width balance before and after test and in part IV we conclude. This application research is aimed to have a positive guiding significance for the AFPM production. We hope that our description of prototype developing process and the solutions we took to deal with the manufacturing problems can have a positive influence for future research.

**References:**

 [1] Edward, Iman, Said Wahsh, and M. A. Badr. "Analysis of PMSM drives for electric vehicles." Proceedings of the 37th SICE Annual Conference. International Session Papers. IEEE, 1998. [2] Situ, Lixin. "Electric vehicle development: the past, present & future." 2009 3rd International Conference on Power Electronics Systems and Applications (PESA). IEEE, 2009. [3] P. Campbell, "Principles of a permanent-magnet axial-field d.c. machine," in Proceedings of the Institution of Electrical Engineers, vol. 121, no. 12, pp. 1489-1494, December 1974. [4] P. Campbell, “The magnetic circuit of an axial-field DC electrical machine”, IEEE Transactions on Magnetics, Vol.Mag11, No.5, Sept. 1975, pp.1541-1543. [5] E. Spooner and B. J. Chalmers, “TORUS, a slotless, toroidal stator, permanent magnet generator”, Proc. IEE, Part-B, Vol.139, No. 6, Nov. 1992, pp. 497-506. [6] F. Caricchi, F. Crescimbini and E. Santini, “Basic principles and design criteria of axial-flux PM machines having counterrotating rotors”, IEEE Transactions on Industry Applications, Vol.31, No. 5, Sept/Oct 1995, pp.1062-1068. [7] Aydin. M, Huang. S and Lipo, T.A. (2004). Axial flux permanent magnet disc machines: A review. Conf. Record of SPEEDAM. [8] Akatsu, K. and Wakui, S. , “A comparison between axial and radial flux PM motor by optimum design method from the required output NT characteristics”, in international journal for computation and mathematics in electrical and electronic engineering, Vol. 25 No. 2, pp. 496-509.

**Images:**

![](https://assets.underline.io/uploads/markdown_image/1/image/8d70807489b42e8db65cc0ef3398b370.jpg)
Impact of PM Segment Number on Rotor PM Eddy Current Loss

![](https://assets.underline.io/uploads/markdown_image/1/image/d5d466e7ccd06071dfca97e53692f578.jpg)
Rotor magnetic flux density distribution

 Axial-flux Permanent Magnet Machine Application Research with Strap-wire Technique and Difference Analysis of Rotor Eddy-current Loss between Simulation and Test

I. INTRODUCTION With the increasingly serious problems of energy shortage and environmental pollution, the development of electric vehicles has become the focus of the world. Electric motors and their drive systems are one of the largest electrical energy consumptions, consuming more than twice as much as lighting, the next largest end-use, accounting for about between 43% and 46% of all global electricity consumption [1]. Therefore, it is necessary to further improve the overall efficiency of electric motors, especially for New European Driving Cycle (NEDC). More and more scholars improve motor efficiency by changing motor materials [2-6]. Grain-Oriented electrical steel materials (GO materials) are mainly used in the core manufacturing of transformers, reactors and large generators. In recent years, GO materials are gradually introduced into motors. Compared with the non-oriented electrical steel materials (NO materials) commonly used in motors, the GO material has the advantages of lower core loss characteristic and higher saturation magnetic flux density. However, the GO material has different permeability and core loss in different directions, so it is difficult to design an axial-flux permanent magnet (AFPM) motor with yoke to take advantage of the GO material. The main flux of the AFPM motor with yokeless stator studied in this paper only passes through the stator in one direction, eliminating the disadvantages of AFPM motor with stator yoke. This paper discusses the application of GO materials in AFPM motors with yokeless stator, compares the basic characteristics of the GO material and NO material used in AFPM motor, analyses the difference between two axial-flux topologies in the use of materials and verifies through finite-element analysis (FEA). II. DESCRIPTION OF THE AFPM MACHINE AND GRAIN-ORIENTED ELECTRICAL STEEL SHEET The GO material has electromagnetic anisotropy: the electromagnetic properties in different directions are different, as is shown in Fig.1(a). In the easy magnetization direction of GO material, it can be seen from Fig.1(b-e) that it is easier to magnetize than NO material, with higher saturation point and lower core loss. However, in the direction of hard magnetization, the GO material is harder to magnetize and has higher core loss. So reasonable use of the easy magnetization direction can provide better torque output ability and less core loss for motors. The explosive view of the AFPM machine investigated in the paper with 48-slot /8-pole is shown in Fig.2, which belongs to the axial-flux topology with a single-stator double-rotor. Main parameters of the motor are shown in Tab. I. Due to the arrangement of the PMs, the main magnetic flux paths only pass through the stator in one direction, which can be set as easy magnetization direction, without passing through the stator yoke tangentially, as shown in Fig.3. III. FEA AND PERFORMANCE Due to AFPM machine’s particular 3-D configuration, 3-D FEA is carried out. Fig.4 shows the stator flux distributions and flux density under no load, showing that the main flux paths go through the easy magnetization direction. It should be noted that even if the GO material is easier to magnetize, the air-gap flux density and back-EMF are not greatly improved under no-load condition due to the large air-gap and PM reluctance. With the increase of armature current, the advantages of GO material are gradually manifested. The torque output capacity of the AFPM machine with GO material and NO material obtained by 3-D FEA is shown in Fig.5, reflecting that the motor with GO material has better torque output capacity than that of NO material. Taking Irms=530A as an example, the torque of GO material is about 15 N.m higher than that of NO material, increased by about 3%. Fig.6 shows the efficiency map of the motor with different materials. The GO material not only improves the torque, but also improves the efficiency. Tab. II compares torque, core loss and efficiency of motors with yokeless stator using different materials under typical working conditions. Similarly, this paper also tried the application of GO material in AFPM machine with stator yoke. As is shown in Fig.7, since a tangential magnetic circuit passes the stator yoke, it is hard to avoid the different magnetization direction when using the easy magnetization direction. Tab. III compares the torque and efficiency of the AFPM machine with yoke under several working conditions. With the stator using GO material, the torque output capacity is lower than that with NO material, and core loss is also increased. Hence, it is not recommended to use GO material in this AFPM topology. IV. CONCLUSION By introducing GO material to yokeless stator, an attempt is made in this paper to use its electromagnetic anisotropy to make the main magnetic flux paths of the motor passing through its easy magnetization direction, without crossing hard magnetization direction, so as to improve torque performance and reduce core loss as well as efficiency. Base on the 3-D FEA model of 48-slot /8-pole AFPM machine, the field distributions, flux density, torque output capacity and efficiency map have been compared between GO and NO material. Benefitting from GO material and yokeless structure, the torque output capacity of AFPM machine is increased by about 5%, the core loss is also reduced by 10%, and the efficiency is improved by 2%. Furthermore, the inapplicability of GO material in the AFPM with stator yoke is also discussed and verified in this paper.

**References:**

[1] P. Waide and C. U. Brunner, “Energy-efficiency policy opportunities for electric motor-driven systems,” International Energy Agency (IEA), 2011. [2] W. Geng, Z. Zhang and Q. Li, "High Torque Density Fractional-Slot Concentrated-Winding Axial-Flux Permanent-Magnet Machine With Modular SMC Stator," IEEE Transactions on Industry Applications, vol. 56, no. 4, pp. 3691-3699, July-Aug. 2020. [3] D. Kowal, P. Sergeant, L. Dupre and A. Van den Bossche, "Comparison of Nonoriented and Grain-Oriented Material in an Axial Flux Permanent-Magnet Machine," IEEE Transactions on Magnetics, vol. 46, no. 2, pp. 279-285, Feb. 2010. [4] J. Ma et al., "Optimal Design of an Axial-Flux Switched Reluctance Motor With Grain-Oriented Electrical Steel," IEEE Transactions on Industry Applications, vol. 53, no. 6, pp. 5327-5337, Nov.-Dec. 2017. [5] Y. Tsuchiya and K. Akatsu, "A Study of the Switched Reluctance Motor using Grain-Oriented Electrical Steel Sheets," 2020 IEEE Energy Conversion Congress and Exposition (ECCE), Detroit, MI, USA, 2020, pp. 3623-3628. [6] S. Javadi, M. Mirsalim and J. Sabzivand, "3-D FEM analysis of a novel structure for axial-flux homopolar generators using a grain-oriented silicon steel stator core," 2008 International Conference on Electrical Machines and Systems, Wuhan, 2008, pp. 3640-3643.

**Images:**

![](https://assets.underline.io/uploads/markdown_image/1/image/2e470724e420eb5e24f1d276f4bcb0de.jpg)

![](https://assets.underline.io/uploads/markdown_image/1/image/9c74136a3b306c49ec4636f896a67e57.jpg)

 Electromagnetic Analysis and Efficiency Improvement of Axial-Flux Permanent Magnet Motor with Yokeless Stator by using Grain-oriented Silicon Steel Material

I. INTRODUCTION Due to the absence of neither permanent magnets (PMs) nor windings on the rotor, stator wound-field synchronous (SWFS) machines [1]-[5] were presented for potential applications to domestic appliances and more-electric aircraft, and renewable energy, etc. SWFS machines have the advantages of the eliminations of brushes and slip rings, low cost, fault tolerance, simple as well as a robust double salient structure similar to switched reluctance machines (SRMs) [6], which are suitable for critical-safety applications. Besides, the demerits of high torque ripple, acoustic noise, and vibration in SRM are alleviated with the SWFS machines which can be fed by a standard three-phase full-bridge inverter. Nevertheless, the existence of additional DC windings makes SWFS machines suffer from restricted torque density and redundant DC power source. Thus, the variable flux reluctance machines (VFRMs) [5] [7] with integrated DC bias and AC current excitations were developed by using an open-winding winding configuration to offer DC excitations. However, the inherently unipolar coil flux linkage in VFRMs still results in a relatively low torque density. Therefore, this paper attempts to propose a novel wound field switched flux (WFSF) machine with a zero-sequence field current excitation. An open-winding circuit is employed to generate a zero-sequence current to offer DC excitation. Moreover, the SF working principle will be employed to improve the torque capability compared to the VFRM. In this paper, the machine topology, operating principle, and design considerations will be described, respectively. Then the major electromagnetic performance will be analyzed and compared with the conventional WFSF counterpart, which confirms the superiority of the proposed design. II. MACHINE TOPOLOGIES AND OPERATING PRINCIPLE Figs. 1 (a) and (b) show the WFSF machines with conventional separate and integrated DC and AC excitations. It demonstrates that the two machines are with identical stator/rotor doubly salient structure and single-layer overlapping winding layout. The main difference lies in the winding functions of the two sets of windings in the two WFSF machines. In the conventional WFSF machine [2]-[4], one set of winding serves as continuous DC excitation while another works as the armature winding. On the other hand, two sets of windings are both injected with integrated AC plus DC bias currents in the WFSF machine with zero-sequence field current excitation. As a result, more copper area is available for the armature windings in the proposed WFSF machine with integrated “AC+DC” current excitations. The control circuit is illustrated in Fig. 1(c), in which the open-winding-based dual three-phase inverter with a single voltage source is utilized [7], and the output terminals of two inverters are connected to both ends of the stator windings. This method is the utilization of the constant zero-sequence current acting as the field current, while two three-phase currents are controlled by two inverters. The operating principle of the proposed WFSF machine is shown in Fig. 1(d). For each group armature windings, such as A1 and B2, when A1 is injected with continuous DC current, and the sinusoidal flux linkage will be induced in B2, and vice versa. As a result, when an AC sinusoidal current with a DC bias is applied to both two coils A1 and B2, the effective electromagnetic energy conversion will be achieved similar to those conventional WFSF counterparts [2]-[4]. III. ELECTROMAGNETIC PERFORMANCE AND COMPARISON Fig. 2(a) shows the open-circuit coil and phase flux linkage waveforms, it can be the DC biases in two individual coils are canceled, and a bipolar and sinusoidal phase flux linkage can be obtained. Besides, the open-circuit phase back-EMF waveforms at 400 r/min are shown in Fig. 2(b). It can be observed that the back-EMF is symmetric and sinusoidal in the WSFC machine with zero-sequence field excitation, while asymmetric non-sinusoidal for the conventional case. This is mainly attributed to the fact that the dual three-phase winding arrangement in the proposed machine allows the even-order harmonic cancellation within one phase. Besides, the steady torque and torque versus current characteristics of the two machines are compared as shown in Figs. (c) and (d), respectively. It should be noted that the winding turns of the two machines are designed as the same for a fair comparison. As a result, it demonstrates the WSFC machine with zero-sequence field excitation exhibits 68% higher magnitudes of steady torque, together with 30.4% lower torque ripple than the conventional case. Meanwhile, it demonstrates that the average torques of the proposed machine exceed that of the conventional counterpart regardless of the armature current excitations. Thus, better electromagnetic performance can be obtained with the proposed design with zero sequence field excitation. The detailed characteristics analysis, comparison results, and experimental verification will be presented in the full paper.

**References:**

[1] Z. Q. Zhu, Y. J. Zhou, J. T. Chen and J. E. Green, “Investigation of nonoverlapping stator wound-field synchronous machines.” IEEE Trans. Energy Convers., vol. 30 no. 4, pp. 1420-1427, May. 2015. [2] H. Pollock, C. Pollock, R. T. Walter, and B. V. Gorti, “Low cost, high power density, flux switching machines, and drives for power tools,” in Conf. Rec. 38th IEEE IAS Annu. Meeting, Oct. 2003, vol. 3, pp. 1451–1457. [3] Y. Tang, J. J. H. Paulides, T. E. Motoasca, and E. A. Lomonova, “Flux switching machine with DC excitation,” IEEE Trans. Magn., vol. 48, no. 11, pp. 3583-3585, Nov. 2012. [4] B. Gaussens, E. Hoang, O. Barriere, J. Saint-Michel, and M. Gabsi, “Analytical approach for air-gap modeling of field-excited flux-switching machine: No-load operation,” IEEE Trans. Magn., vol. 48, no. 9, pp. 2505–2517, Sep. 2012. [5] X. Liu and Z. Q. Zhu, “Electromagnetic performance of novel variable flux reluctance machines with dc-field coil in stator,” IEEE Trans. Magn., vol. 49, no. 6, pp. 3020-3028, Jun. 2013. [6] K. Vijayakumar, R. Karthikeyan, S. Paramasivam, R. Arumugam, and K. N. Srinivas, “Switched reluctance motor modeling, design, simulation, and analysis: a comprehensive review,” IEEE Trans. Magn., vol. 44, no. 12, pp. 4605-4617, Dec. 2008. [7] Z. Q. Zhu, B. Lee, and X. Liu, “Integrated field and armature current control strategy for variable flux reluctance machine using open winding,” IEEE Trans. Ind. Appl., vol. 52, no. 2, pp. 1519-1529, Mar./Apr. 2016.

**Images:**

![](https://assets.underline.io/uploads/markdown_image/1/image/16bc6a2168fbe73c6a5b2b933b4fc140.jpg)
Fig. 1. Topologies of the WFSF machines (a) with zero-sequence field current excitation and (b) conventional separate AC and DC windings. (c) Control circuit. (d) Operating principle.

![](https://assets.underline.io/uploads/markdown_image/1/image/ac8f544001c98b644d34bbc569f7b5d0.jpg)
Fig. 2. Electromagnetic characteristics comparison. (a) Flux linkages in the proposed machine. (b) Back-EMFs. (c) Steady-state torque. (d). Torque versus current characteristics.

 A Novel Wound Field Switched Flux Machine with Zero-Sequence Field Current Excitation

Wound rotor synchronous machines are widely used because of their advantages, such as simple control, cost efficiency, and stable operation [1]. Because of these advantages, the wound rotor synchronous generator (WRSG) is used in large factories, aircraft, ships, and hydroelectric power generation that require a large power supply [2]. The WRSG does not have a rare earth magnet, and it forms a magnetic field through the rotor winding, as shown in Fig. 1(a). A magnetic field is formed by applying current to the rotor windings through an exciter, and the magnetic flux density can be controlled using the exciter to increase the power density. The core of many electrical machines is made of electrical steel with nonlinear characteristics, and the magnetic flux density must be adjusted to a level where the core is unsaturated to satisfy the user’s desired output power, efficiency, and voltage. Therefore, it is necessary to evaluate the electromagnetic characteristics through electromagnetic analysis during the initial design. Finite element analysis (FEA) software is commercially available because it can obtain accurate analysis results that reflect nonlinearity. However, FEA consumes significant analysis time in the initial design, in which the design parameters change frequently [3]. Therefore, in this study, we propose a nonlinear magnetic equivalent circuit (MEC) to predict the electromagnetic and power generation characteristics of the WRSG. The magnetic equivalent area was divided according to the magnetic path of the WRSM (Fig. 1(b)). The paths through which the magnetic flux passes are divided into stator, rotor, air-gap, and interpolar regions. The magnetic flux inside the circuit could be calculated using the magnetomotive force(MMF) and reluctance, and the flux density in air gap could be calculated using the magnetic flux. Fig. 2(a) shows a comparison of the air gap flux density derived through the MEC and FEA when a field current is applied. The air flux density derived for the MEC represented the flux density waveform when the rotor position rotated, which could be used to derive the stator flux linkage waveform according to the rotor position. The derived flux linkage induced a back electromotive force (EMF) at no load (Fig. 2(b)). In the initial position, the rotor was positioned such that the flux linkage to the windings of one phase was maximized, and the back EMF through the flux that varied according to the rotation of one slot pitch was derived. The back EMF was influenced by the MMF of the rotor winding, and beyond the linearly increasing region, it became saturated because of the nonlinearity of the core material. The back EMF derived from the nonlinear MEC increased linearly and then nonlinearly, owing to saturation beyond a certain field current (Fig. 2(c)). The back EMF obtained using the nonlinear MEC was compared with those obtained through FEA and experimental measurements; the comparison validates the nonlinear MEC proposed in this study. It is necessary to determine the values of the back EMF, inductance, and phase resistance, which are equivalent circuit parameters, to predict the power generation characteristics of a WRSG. A MEC based on the rotor position must be constructed to reflect the inductance that varies with the rotor position, owing to the salient structure of the WRSG. Fig. 2(d) shows the prediction results of power generation characteristics using the inductance derived through the MEC, back EMF, and phase resistance. The FEA results indicated the power generation characteristics in the low-voltage regulation rate region, and the MEC results were in good agreement with the FEA results. Even when the rotational speed of the WRSG changed, the MEC results were in good agreement with the FEA results. By predicting power generation characteristics, the speed and power required by the designer can be predicted, which is useful in the initial design. The nonlinear MEC and analysis results of the WRSG will be presented in further detail in the main paper.

**References:**

[1] S. Nuzzo, M. Degano, M. Galea, et al, IEEE Trans. Ind. Electron., vol. 64, no. 3, pp. 1958–1970, Mar. 2017. [2] A. Griffo, D. Drury, T. Sawata, et al, IEEE Trans. Ind. Electron, vol. 59, no. 9, pp. 3579–3587, Sep. 2012. [3] H.W. derbas, J.M. Williams, A.C. Koenig, et al, IEEE Trans. Energy Convers., vol. 24, no. 2, pp. 388–396, Jun. 2009.

**Images:**

![](https://assets.underline.io/uploads/markdown_image/1/image/30110081fcfd5756909893a64d2ecd83.jpg)
Fig 1. (a) WRSG analysis model and rotor core experimental model; (b) Construction of nonlinear MEC

![](https://assets.underline.io/uploads/markdown_image/1/image/0abfb2acf0d58abdf081b148aaeb8928.jpg)
Fig 2. Comparison of electromagnetic characteristics obtained using the FEA, MEC, and experimental methods: (a) Comparison of air-gap flux density obtained using FEA and MEC methods; (b) Waveform of back EMF at rated speed of 900 rpm; (c) Variation of back EMF with field current; (d) Power-voltage curve

 Prediction of power generation performance of wound rotor synchronous generator using nonlinear magnetic equivalent circuit method

Compared with the conventional motors used on the ground, the DC starter-generators (DCSGs) for aero-engines have higher electrical, thermal, magnetic and mechanical loads. Excessive electrical loads will increase the armature response of the (DCSGs) and worsen the commutation, thus shortening the lifetime and reducing the reliability of the (DCSGs). In order to improve the lifetime and reliability of the (DCSGs), this paper studied the effects of commutator structure, compensation windings, commutating poles and brush sizes on commutation effect of DCSGs based theory analysis combined with numerical analysis. Then effective measures to improve the commutation performance of this type of DCSGs are proposed, and the commutation spark observation test of a certain type of DCSGs verifies the effectiveness of the optimization design measures proposed in this paper. Index Terms：DC starter-generators (DCSGs); Brush; Commutation; Optimal design. I. Introduction The DC starter-generators (DCSGs) used in aero-engine is installed on the engine accessory box. When the aero-engine starts, it acts as a starter motor to drive the engine to ignite. After the engine ignites, it acts as a generator to supply power to the aircraft network. Due to the special application environment of the DCSG, it has the characteristics of low voltage and high current compared with the motor used on the ground. The commutation process is extremely bad, and it may produce larger commutation sparks or even ring fire, shortening the lifetime of the brush and reducing the operational reliability of DCSGs[1][2][3][4]. This paper optimizes the design of parameters such as commutator structure, commutating pole winding and brush size, and studies the influence of parameter changes on the commutation performance of DCSGs under the conditions of low voltage and high current, and discusses the effectiveness of improving the commutation performance of this type of DCSGs The method provides a reference for commutation spark suppression in commutation of the same type of DCSGs. II. Large Current Commutation Modeling The electromotive force and current in the armature winding of the DCSG form an AC-DC voltage and current conversion through the cooperation of the commutator and the brush. This conversion process is called commutation. There is a complicated electromagnetic process in the commutation process, and reactance electromotive force is generated in the commutation element. By reducing the armature reaction and adjusting the brush width to reduce the reactive electromotive force, the commutation spark can be reduced and the commutation performance can be improved. III. Optimal Design Take a 9kW, 6250rpm～12000rpm DCSG as an example to study the influence of commutator structure selection, compensation winding, commutating pole and brush size on DCSG commutation performance. Finite element analysis model of the DCSG is shown in Fig.1. IV. Experimental Verification A start-up test was carried out on the simulated engine load characteristics of a 9kW a DCSG. The test bench is shown in Fig.2. No visible commutation sparks were observed in the test, which verified that the effectiveness of the commutation performance optimization design method proposed in this paper. V. Conclusion This paper studies the optimization design of DCSG to improve its commutation performance. Based theory analysis combined with numerical analysis, the influence of different commutator types, compensation windings, commutating pole windings and brush parameters on the commutation performance of the DCSG is analyzed. The prototype is tested for starting and generating performance. The experimental results verify the effectiveness of the optimized design method in improving the commutation performance, which can provide a reference for improving the commutation performance of the same type of DCSG, and is useful for improving the operating reliability and lifetime of the DCSGs.

**References:**

[1] Takashi, Fukutsuka and Yasunobu. “Arc discharge during commutation process in DC motor immersed in ethanol: approach to contact state of brush on commutator segment”, IEEJ Transactions on Power and Energy, Vol. 139, pp.293-301,2019. [2] H. Kim, H. Kim and W. Moon. “Design optimization of brush permanent magnet DC motors using response surface method”, Journal of Magnetics, vol. 25, pp. 355-362, 2020. [3] K. Sawa, M. Isato and T. Ueno. "Commutation characteristics and brush wear of DC motor at high rotation speed," 2017 IEEE Holm Conference on Electrical Contacts, pp. 176-181, 2017. [4] Z. Qingliang, Takahiro and Ueno. “The new commutation theory to the optimization technique of the large DC machine commutation design”, IEEJ Transactions on Industry Applications, Vol. 131, pp. 654-666, 2011.

**Images:**

![](https://assets.underline.io/uploads/markdown_image/1/image/246f78fbf529af15431c6c92f934689e.jpg)
Fig.1. Finite element analysis model of the DCSG.

![](https://assets.underline.io/uploads/markdown_image/1/image/acbe807bc1de68577c1c67ffac5a574f.jpg)
Fig.2. Test bench of DCSG.

 Research on Optimal Design of Commutation Performance of Starter-Generator Used in Aero-Engine

I.Introduction Permanent magnet synchronous machine (PMSM) is widely recognized as a competent candidate for automotive propulsion [1], which offer the merits of high power density and high efficiency. Nevertheless, due to the inverter voltage and current limitations, the air-gap flux of the conventional PMSM is difficult to adjust even though a continuous negative d-axis current is injected for flux-weakening under high-speed operation [2]. As a result, how to realize effective air-gap flux adjustment, and further extend the speed range as well as high efficiency operation is regarded as a major challenge for the conventional PMSMs. Fortunately, the concept of variable flux memory machine (VFMM) is proposed to address the above issues [3]-[4]. The low coercive force (LCF) PM can be varied via a magnetizing or demagnetizing current pulse, in which the air-gap flux density can be flexibly adjusted. Generally, high coercive force (HCF) and LCF PMs are simultaneously employed in VFMM to realize high torque density and excellent flux adjustability [5]-[6]. Due to the magnetization state (MS) stabilization effect of HCF PMs on LCF PMs in series magnetic circuit VFMM, the unintentional on-load demagnetization issue can be avoided. Nevertheless, it also brings the problems of limited flux regulation range and higher required current pulse level for full remagnetization, which are undesirable for practical applications [7]. Therefore, this paper proposes a novel asymmetric-PM VFMM (APM-VFMM) with anti-demagnetization barrier design to realize wide flux regulation range and excellent on-load demagnetization withstand capability. The topology and features of the proposed APM-VFMM are introduced, respectively. The electromagnetic characteristics under different MSs are evaluated and compared. The efficiency maps are investigated to verify the advantages of the proposed design in terms of excellent flux regulation range and high efficiency operation over a wide speed range. II.Machine Configuration The topology evolution of the proposed APM-VFMM is shown in Fig. 1, in which the magnetic barrier is designed to resist the on-load unintentional demagnetization of the LCF magnet. The “U”-shaped magnetic barrier is added to the conventional series magnetic circuit VFMM, thereby forming into the anti-demagnetization barrier design. Then, the LCF magnet can be a substituted for the HCF magnet on one sided portion of the barrier design to form an asymmetric-PM configuration. The corresponding cross section of the proposed machine is shown in Fig. 2, which are characterized by a dual-layer PM structure [7]. The “U”-shaped magnetic barrier of the first PM layer can be employed to protect the working point stability of LCF PMs. For the second PM layer, asymmetric spoke-type hybrid PM arrangement is located on the d-axis, which can reduce magnetic saturation and enlarge flux regulation range. Each LCF magnet has an adjacent spoke-type HCF magnet to form a series magnetic circuit, achieving excellent demagnetization withstand capability. III.Electromagnetic Performance The open-circuit air-gap flux density waveforms of the proposed APM-VFMM under different MSs are plotted in Fig. 2(a) and (b), in which the flux-enhanced and flux-weakened states refer to the cases that the LCF magnets forward or reverse magnetized by twice the rated current considering inverter capacity, respectively. It demonstrates that the magnetic field in the flux-weakened state is significantly lower than flux-enhanced state, which implies that an expected flux-weakening effect can be achieved. Furthermore, the torque characteristics of the proposed machine are plotted in Fig. 2(c) and (d). It can be observed that the proposed machine in flux-enhanced state provides higher output torque than that of the flux-weakened state. Finally, the efficiency maps of the proposed machine under different MSs are shown in Fig. 2(e) and (f), which confirm that the proposed machine can achieve high efficiency maintaining by selecting appropriate MS under different operating regions.

**References:**

[1] I. Boldea, L. N. Tutelea, L. Parsa, and D. Dorrell, IEEE Trans. Ind. Electron., vol. 61, no. 10,pp. 5696-5711, Oct.2014. [2] W. L. Soong and T. J. E. Miller, IEE Proc. Electr. Power Appl., vol. 141, no. 6, pp. 331-340, Nov. 1994. [3] V. Ostovic, IEEE Ind. Appl. Mag., vol. 9, no. 1, pp. 52-61, Jan./Feb. 2003. [4] H. Hua, Z. .Q. Zhu, A. Pride, R. P. Deodhar, and T. Sasaki, IEEE Trans. Ind. Appl., vol. 53, no. 5, pp. 4396-4405, Sept./Oct. 2017. [5] A. Athavale, K. Sasaki, B. S. Gagas, T. Kato, and R. D. Lorenz, IEEE Trans. Ind. Appl., vol. 53, no. 5, pp. 4318- 4326, Sep./Oct. 2017. [6] H. Yang, S. Lyu, H. Lin, Z. Q. Zhu, H. Zheng, and T. Wang, IEEE Trans. Ind. Electron., vol. 67, no. 7, pp. 5258-5268, July 2020. [7] H. Yang, H. Lin, Z. Q. Zhu, S. Lyu, and H. Wang, in Proc. Energy Convers. Congre. Expo (ECCE 2018), Portland, OR, Sept. 23-27, 2018.

**Images:**

![](https://assets.underline.io/uploads/markdown_image/1/image/de9377d8916ba31d300acd96c34980c4.jpg)
Fig. 1. Machine topology. (a) Topology evolution. (b) Cross section of the proposed APM-VFMM.

![](https://assets.underline.io/uploads/markdown_image/1/image/bf10326bea48e5d7b69487091f164d07.jpg)
Fig. 2. Electromagnetic characteristics. (a) Open-circuit air-gap radial flux densities. (b) Harmonic spectra. (c) Steady-state torque waveforms. (d) Torque-angle waveforms. (e) Efficiency maps under flux-enhanced state. (b) Efficiency maps under Flux-weakened state.

 A Novel Asymmetric-PM Variable Flux Memory Machine with Anti-Demagnetization Barrier Design

Introduction: The torque ripple can hardly be avoided for synchronous reluctance motors (SynRMs) in load conditions. It is generated by the interaction of the stator and rotor electromagnetic field harmonics, which have the same number of pole pairs but different rotating angular frequencies. In the air gap, the flux density is equal to the magnetomotive force (MMF) times the permeance. In [1], the air-gap field was analyzed, and a method for designing low-torque-ripple SynRMs was proposed. However, it did not consider the rotor permeance distribution and was limited to a relatively symmetrical rotor topology. In [2], it was verified that the torque ripple is highly affected by the position of the flux-barrier ends. The introduction of non-uniform flux barriers is capable of reducing the torque ripple. Based on the recent air-gap field modulation theory [3], the SynRM rotor with multi-layer flux guides, or flux barriers, can modulate extra harmonics in the air gap, whose frequencies are different from the primitive MMF. It is considered that the topology of the SynRM rotor is usually asymmetric, especially when a rotor with non-uniform flux barriers is adopted, and a series of permeance harmonics are produced. However, it is quite difficult to calculate the rotor permeance by analytical methods when the flux-barrier ends distribute along the rotor surface non-uniformly. This study proposes a method to analyze the effects that the rotor topology has on the rotor permeance harmonics by a virtual permanent magnet harmonic machine (VPMHM). By adjusting the rotor topology, certain permeance harmonics resulting in tremendous torque ripple can be avoided. Instead of simulating the output torque waveform directly, the proposed method can accelerate the optimization process and save much time. Analysis model: In [4], the concept of a VPMHM was introduced to generate harmonics with sinusoidal magnetization. Fig. 1(a) shows the model of the VPMHM, and Fig. 1(b) shows the implementation of the VPMHM in the optimization process for SynRMs in this study. The VPMHM has two parts, an ideal stator yoke whose permeability is infinite and an annular permanent magnet (PM) with the sinusoidal magnetization. The VPMHM is capable of generating sinusoidal distributed flux density in the air gap. Once an actual rotor is put in, the flux density waveform is no longer the pure sinusoid because of the modulation of the flux guides. The harmonics can be regarded as the effects of the rotor permeance. In this article, a 36-slot, 4-pole SynRM model is taken as an example to investigate the effects of the rotor permeance and perform the finite-element based optimization. The rated phase current is 130 Arms, and the motor is controlled by the maximum torque per ampere (MTPA) strategy. Results: An initial rotor was fast established based on some specific rules. The rotor permeance is mainly determined by the positions of the flux-barrier ends. With the help of the VPMHM, there is no need to analyze the entire dynamic process. A large-scale enumeration calculation is carried out for the air-gap electromagnetic field under different positions of the flux-barrier ends. The rotor that can create the minimum harmonics, whose orders are the same as the stator harmonics is preferred. Compared with the absolute values, per unit values are used to describe the magnitude which makes the comparison easier and more straightforward during the optimization process. Fig. 2(a) shows the fast Fourier transform (FFT) results of the air-gap flux density harmonics before and after the optimization. Though both of them contain considerable 21st and 23rd order harmonics, the effects on torque ripple are minor because the magnitudes of the corresponding stator harmonics are relatively small. The 17th and 19th order harmonics are evidently suppressed, which have the same number of pole pairs as the stator slot harmonics. Hence, the torque ripple caused by the slot harmonics can be mitigated. The fifth and seventh order harmonics slightly increase, but the influences of them are not severe as that of the stator slot harmonics when a short-pitched double-layer winding is adopted. For validation, Fig. 2(b) shows the results of the output torque waveform as a function of time at the nominal load. Compared with the initial rotor, it is salient that the torque ripple is effectively suppressed after optimization.

**References:**

[1] A. Vagati, M. Pastorelli, G. Francheschini, IEEE Transactions on Industry Applications, vol. 34, no. 4, pp. 758-765, July-Aug. 1998, doi: 10.1109/28.703969. [2] N. Bianchi, S. Bolognani, D. Bon and M. Dai Pre, IEEE Transactions on Industry Applications, vol. 45, no. 3, pp. 921-928, May-june 2009, doi: 10.1109/TIA.2009.2018960. [3] M. Cheng, P. Han and W. Hua, IEEE Transactions on Industrial Electronics, vol. 64, no. 8, pp. 6063-6074, Aug. 2017, doi: 10.1109/TIE.2017.2682792. [4] C. Di, I. Petrov and J. J. Pyrhönen, IEEE Transactions on Magnetics, vol. 54, no. 12, pp. 1-12, Dec. 2018, Art no. 8111012, doi: 10.1109/TMAG.2018.2873279.

**Images:**

![](https://assets.underline.io/uploads/markdown_image/1/image/945e93b891cd547a409babe20dea51a8.jpg)
Fig. 1 Concept of the VPMHM and the flux density distribution in the air gap. (a) Model of the VPMHM (b) Implementation of the VPMHM with the actual rotor.

![](https://assets.underline.io/uploads/markdown_image/1/image/4c46c91241441ae7dee9138088d02dd8.jpg)
Fig. 2 The results before and after the optimzation. (a) Harmonics produced by the SynRM rotor under the excitation of a VPMHM. (b) The output torque waveform as a function of time at the nominal load.

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